Picoammeter Design

[bsw_m]

As for ADA4530-1. This is an excellent opamp, suitable for making a device that will be able to measure current up to 1E-16A, albeit with some nuances.

[SilverSolder]

An abacus?   

We can say that the abacus for individual electrons

I just read your other thread about that instrument.  Those are some pretty cool toys you've got there! 

[David Hess]

The LMP7721 seems to be the only device where they have made the effort to optimise the pinout specifically for low leakage. Although it is only available in SO8, it has two dedicated n/c pins between the inputs and other pins, that can be included in the guarding scheme. The price seems to have come down too (at least looking at RS in the UK) it's about a third of the price of an LMC6001 (probably a result of the additional part selection step in the latter).

They would not have much choice in a surface mount package.  With the DIP LMC6081, there is plenty of space to place guard traces, or the input pin can be bent up to horizontal and air wired.

In the past they had a 10-lead TO-99 through-hole package for low input bias current parts.

[MegaVolt]

1G 1% resistor (from Taobao)

Can you share the link?

[MegaVolt]

It is possible instead of a large resistor to use a T-shaped divider of 3 resistors. That will allow to apply smaller resistors and reduce the noise level.

[Kleinstein]

It is possible instead of a large resistor to use a T-shaped divider of 3 resistors. That will allow to apply smaller resistors and reduce the noise level.

One can use the T network to replace a large resistor. However as a TIA / pA meter this comes with a higher noise level. For a TIA the input referred current noise goes down with a larger resistor. The larger resistor will have more voltage noise, going up with the square root of the resistance. However the current noise is voltage noise divided by the resistor and thus proportional to 1 over the square root of the resistance.

For a classical TIA with a resistor feedback, the noise of the resistor can be a major contribution and the very high level resistors (> 100Mohms) often have comparatively poor stability. For the very low current range there is the alternative to use a charge amplifier instead and than look at the rate of ouput voltage change. Good capacitors in the 100 pF range are better available than good resistors in the Gohms range. Another, a bit unusual way is to use a photodiode as feedback element - this gives some extra shot noise, but can still be a viable alternative.

[magic]

The short answer is that noise of R2 decreases, but it is now amplified by the ratio of R2/R1.
It's same thing as using a lower resistor and adding a noninverting gain stage.
If you do the math, you see you cannot win.

[MegaVolt]

If you do the math, you see you cannot win.

Good We do not win noise. But can we win in the stability and accuracy of resistors?

[Kleinstein]

The T circuit can have more stable / accurate resistors, as it can use lower value ones.  Above some 10 M it is rare to find really stable ones (e.g. wire wound).

[magic]

The T network also amplifies the opamp's flicker noise while preserving the original signal gain.
You throw away SNR and there will be a point where no amount of filtering will recover it.

By all means go and calculate the details or build a prototype if you want.

[Marco]

So would the lowest leakage opamp you can realistically make without a fab be a composite opamp with a 2n7002 differential pair? (the class 0 ESD ones) Ignoring the terrible offset drift for a moment, oven can take care of that.

[Cerebus]

You'd probably have better luck with a different discrete unprotected IGFET.

First off find something (if you can, choices in discrete small IGFETs are pretty slim nowadays) with a hermetic case - a glass seal will perform better leakage wise than epoxy. Secondly, find something with the substrate brought out to a terminal; then you can mess around with bootstrapping the substrate independently of the source (within limits obviously, you still need to keep any parasitic junctions turned off) and might shave a few percent of leakage off that way. Again, good luck with finding a 4 lead IGFET in anything other than NOS nowadays.

[magic]

Or try your luck with depletion mode MOSFETs, as they can be made to work at zero Vgs.
There is still the drain, but it comes from the bottom of the die and from the opposite side of the package (in SOT23 etc).

Or not from the bottom? Are those things vertical or lateral, actually?

[Kleinstein]

Small unprotected MOSFETs are difficult to get. Essentially all the modern one have a zener for protection and this gives quite some leakage.
Chances are some of the low leakage CMOS OP are easier to get than low leakage MOSFETs.

The other alternative is a small JFET, especially MMBF4117 with small drain-source voltage (e.g. some 2-3 V). This could be similar to the input section of the Keithley 617.

[Cerebus]

My own experience would seem to suggest that the x4117 family are in at best a few hundred fA territory rather than 10s or single fA territory. I think one is probably stuck with integrated CMOS solutions when aiming for absolute minimum leakage/bias current.

[Kleinstein]

The x4117 would not beat the dedicated low bias CMOS chips. In pA meter design the OPs bias is only one hurdle to take. In many designs the feedback resistor and PCB/relay leakage can be the larger problem.  So a low bias OP is onely a part of the solution and with relatively afordable CMOS OPs (e.g. LMC662 or LMC6482), maybe even more like the simpler part.

[Marco]

You'd probably have better luck with a different discrete unprotected IGFET.

Yeah, they seem to have completely ditched all the old parts. Nexperia has an application note from a mere 3 years ago where they still said BSS138P is unprotected ... yet it's AEC-101 certified now.

Linear Systems at least still selling new unprotected small signal MOSFETs, 5$ for a single in a nice TO72 isn't that bad. Just pray the distributor was careful parcelling them out.

[bsw_m]

At the moment I am thinking about the concept of a picoammeter on a vibrating reed capacitor.
Why do I need it.
Some time ago, I purchased about 50 pieces of resonant vibrating reed capacitors DRK-2 "New old stock". These capacitors were developed in specialized design engineering bureau (later became an institute MNIPI) of the Minsk Molotov plant by Shuklin A.S.  in the early 60s. The capacitors I purchased were released in 1987. These capacitors were purchased for study and experimentation. I'm wondering what results can be achieved today using these old parts in the front end of a transimpedance amplifier.
At the moment, the stability of the resonant frequency has been assessed. The leakage of the input insulator was also measured. The leakage current of the input insulator was about 2.5E-16A at a test voltage of 100V.

The resonant frequency measurement is given in the attached graph.
I do not like the stability of the resonant frequency, so I will modify these capacitors to get better stability.

What frustrates me, unfortunately, if I publish the full documentation for the resulting picoammeter, no one will be able to repeat it, due to the fact that unobtanium components  will be used in the input circuits. 

If you are interested, then as experiments and new data are obtained, the results can be published. If there is no interest, then I probably will not litter the forum.

[SilverSolder]

At the moment I am thinking about the concept of a picoammeter on a vibrating reed capacitor.
Why do I need it.
Some time ago, I purchased about 50 pieces of resonant vibrating reed capacitors DRK-2 "New old stock". These capacitors were developed by Shuklin A.S. in the early 60s. The capacitors I purchased were released in 1987. These capacitors were purchased for study and experimentation. I'm wondering what results can be achieved today using these old parts in the front end of a transimpedance amplifier.
At the moment, the stability of the resonant frequency has been assessed. The leakage of the input insulator was also measured. The leakage current of the input insulator was about 2.5E-16A at a test voltage of 100V.

The resonant frequency measurement is given in the attached graph.
I do not like the stability of the resonant frequency, so I will modify these capacitors to get better stability.

What frustrates me, unfortunately, if I publish the full documentation for the resulting picoammeter, no one will be able to repeat it, due to the fact that unobtanium components  will be used in the input circuits. 

If you are interested, then as experiments and new data are obtained, the results can be published. If there is no interest, then I probably will not litter the forum.

The project sounds very interesting, it doesn't matter if it is a one-off...  I'd certainly be interested in reading (reeding?) about it!

[ZhuraYuk]

I think when more details and technical ideas will be revealed, the more people will get interested in the project.
Also to make the implementation repeatable it is better to make it more universal when you can easily replace DRK with TI amp and thus get -16 lower measurement range instead of -18. Someone might even recreate DRK itself or MEMS DRK will appear on market soon.

[bsw_m]

I assembled a prototype amplifier on a slightly modified DRK-2.
The amplifier's own noises were tested, not entirely satisfied with the result. When checking noise, the amplifier worked in buffer mode (gain = 1)
But new additional knowledge was obtained on the operation of the DRK-2. Now I'm thinking about the possibility of creating a DIY vibrating reed capacitor.

[Kleinstein]

Te noise (or background) level looks quite high. It looks like part of this may be some AC background and not all noise. It may be worth looking at the FTT of the data.
The final use would likely be mainly for the very low frequency part ( e.g. << 10 Hz) and not so much the faster part. So the actual noise performance may not be that bad.

[Atomillo]

Hello all!

I've recently read this gem of a thread and started pondering and trying to do some simple noise calculations myself. While doing so, some doubts have appeared to me and I would be grateful if someone more knowledgeable could lend me a hand (I'm not creating a new thread because I believe these questions might be relevant to the original design and posterior discussion, but if not please let me know).

The first question is this: isn't the Johnson noise of the 1 Meg resistor connected directly at the input? Doesn't the low noise we achieve with a high feedback resistor get ruined by this? I assume not, since people with a lot of experience in this area have not mentioned it (in the various noise calculations done by Alex Nikitin it isn't taken into account for example), but I don't see how this is so. Is something cancelling the noise of this resistor?

The second question is related to T-networks. I considered the possibility of using them along with a capacitor, to simulate pF and sub-pF capacitors with larger, more accurate ones. But my pre eliminary (and probably erroneous) calculations show that the johnson noise current of the resistors used in the divider would translate directly to input refered current noise. Is this correct?

Many thanks all for this great thread and any help is very much appreciated!

[Kleinstein]

You are right about the T network: it gives additional gain for the ouput, just like a very high resistor, but the noise is the same as with the smaller actual resistor and thus more noise than a true large resistor. With capacitors the noise is not a problem though.

The protection resistor in series to the OP-amp does contribut, just like noise of the OP-amp. However this is usually not much compared to the noise from the FB resistor. The protection resistor becomes relevant when the FB resistor is comparable or smaller. Another case is when the DUT is relatively low resistance.

[Atomillo]

The protection resistor in series to the OP-amp does contribut, just like noise of the OP-amp. However this is usually not much compared to the noise from the FB resistor. The protection resistor becomes relevant when the FB resistor is comparable or smaller. Another case is when the DUT is relatively low resistance.

This is what I don't get. Shouldn't it be the other way (i.e the protection resistor much larger than the feedback resistor)? After all, the Johnson current of the protection resistor is proportional to 1/sqrt(R) too.
Just as a numerical example, with the 1Meg resistor at T=300K, we have a noise of 0.11pA/rt(Hz). Isn't this noise added directly to the input of the circuit? How is it that in the given calculations previously on this thread only the current noise of the 1G feedback resistor is taken into account? Does the op-amp cancel this noise in any way I'm not seeing?

You are right about the T network: it gives additional gain for the ouput, just like a very high resistor, but the noise is the same as with the smaller actual resistor and thus more noise than a true large resistor. With capacitors the noise is not a problem though

I was refering to the noise of the resistors that make the divider at the output of the op-amp. I arrived to the conclusion that it didn't make sense to remove the high value resistor for a capacitor to get rid of the noise if the noise of the lower-valued resistor in the divider was much worse (since their current noise appeared to translate directly to the input).

Many thanks for your help!
Cai