High-Z wideband buffer amp for 50 ohm CRO inputs

[GK]

Maybe a silly idea, but what about the "classic" solution, using the discrete buffer for AC highspeed signal + DC servo for the rest? (Instead of using veeeeery very fast opamps with fbomb cost level) 

Deliberated over already. OPA653 is ~$4USD.

[Yansi]

I think ur wrong. 150ns settling time is useless for a 500MHz scope. Or have I missed something? Also the output slew rate will be your enemy, depending on the required output swing. You wont be able to do even 200MHz sinewave at 1V amplitude, if I calculate right.

[GK]

I think ur wrong. 150ns settling time is useless for a 500MHz scope. Or have I missed something? Also the output slew rate will be your enemy, depending on the required output swing. You wont be able to do even 200MHz sinewave at 1V amplitude, if I calculate right.

What? OPA653 settling time (to 1%) for a 4V output step is less than 8nS. Slew rate is 2675V/us. The large signal bandwidth (for 2Vpp output) is 475MHz.




The second sentence in my opening post: "I'm aiming for a bandwidth of at least 250 MHz. While this is only half the bandwidth of the 7904 mainframe I feel that there is no point going any more exotic as the achievable bandwidth with 10:1 10M probes limited."

But to clarify, yes, technically there are 10M probes with >250MHz bandwidth (the 500MHz Lecroy PP006A that I use at work for example) but they won't do anywhere near that without a ship load of overshoot and ringing of their own using the earth clip lead. For that you need to detach the the probe claw and ground clip lead and slip one of those fiddly ground contact springs (with a ~5mm tail that you solder directly to the ground of your circuit being probed) over the collar of the exposed contact tip.

[tggzzz]

But to clarify, yes, technically there are 10M probes with >250MHz bandwidth (the 500MHz Lecroy PP006A that I use at work for example) but they won't do anywhere near that without a ship load of overshoot and ringing of their own using the earth clip lead. For that you need to detach the the probe claw and ground clip lead and slip one of those fiddly ground contact springs (with a ~5mm tail that you solder directly to the ground of your circuit being probed) over the collar of the exposed contact tip.

The lower the tip capacitance the better, of course. There are halfway houses; see the HP probe tips shown in the pictures halfway down https://entertaininghacks.wordpress.com/2015/04/23/scope-probe-accessory-improves-signal-fidelity/

[Yansi]

I might then have downloaded a datasheet for another device.  Seems right now.

[David Hess]

I have considered doing the same thing for my Tektronix 7904 but I only have the 7A24 vertical amplifiers which limit bandwidth to 350 MHz.  That is only 50 MHz faster than high impedance passive probes on my fastest oscilloscope which supports them directly.  If I did do something like this, I would first just build a x10 FET probe to take advantage of lower input capacitance.

For an alternative way to implement this without the performance limiting x2 gain stage, check out the Tektronix P6202A 500 MHz x10 FET probe design.

Tektronix kept the source termination to drive the 50 ohm oscilloscope input but instead of a gain stage, they used a fixed x5 input attenuator so the x2 attenuation of the source termination yields x10 at the oscilloscope input.  The fixed input attenuator has the benefit of making the probe input more rugged under all conditions and it attenuates the input capacitance although they still did not add input protection.

Their P6201 900 MHz FET probe design works more like what you are trying with a x2 gain stage to make up for source termination but they used a discrete two transistor differential transconductance amplifier.  Burr-Brown made some fast integrated operational transconductance amplifiers which of course now Texas Instruments produces and I wonder how well the OPA860 without its output buffer would work in this application.  I remember when these came out and I got the feeling that Burr-Brown intended them to become another basic building block like the operational amplifier but it never happened.

[GK]

Tektronix kept the source termination to drive the 50 ohm oscilloscope input but instead of a gain stage, they used a fixed x5 input attenuator so the x2 attenuation of the source termination yields x10 at the oscilloscope input.  The fixed input attenuator has the benefit of making the probe input more rugged under all conditions and it attenuates the input capacitance although they still did not add input protection.

The downside of which is 20dB less sensitivity - 1V/div minimum as the high bandwidth vertical plug-ins are limited to 10mV/div. Since I have a gain of 2 and can switch out the attenuator I get either 1V/div or 100mV/div. This is only 160mVpp output from the OPA653 for full scale deflection, which is far from exercising the op-amps full power bandwidth.

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Made a start on loading the PCB. Am only loading one channel to start with for initial testing. The so called SOT23-5 package of the OPA653 is actually quite beefy compared to an ordinary 3-lead SOT23. I guess that goes some way to explaining the much better thermal performance of this package. All caps are NPO/COG.

It's bed time now. With any luck I'll get this channel completed and powered up tomorrow evening.   

[Yansi]

Very nice build!  Please keep us informed 

[David Hess]

The downside of which is 20dB less sensitivity - 1V/div minimum as the high bandwidth vertical plug-ins are limited to 10mV/div. Since I have a gain of 2 and can switch out the attenuator I get either 1V/div or 100mV/div. This is only 160mVpp output from the OPA653 for full scale deflection, which is far from exercising the op-amps full power bandwidth.

You did not say but talked about 4 channels so I assumed you were using 7A24s which go down to 5 mV/div but of course you are right and it would limit your sensitivity even then to 500mV/div.  Are the extra channels for external triggering?

There *is* a way to get 4 channels of 300 MHz performance from standard passive x10 high-Z probes out of a 7904 with probe tip sensitivities from 200 mV/div to 5 V/div without building anything although it is not very common: the double wide 7A42 logic trigger amplifier plug-in.

[GK]

My 7904 is currently equipped with a 7A19 and a 7A24, giving me three channels:

https://www.eevblog.com/forum/projects/old-school-tek-porn/msg419791/#msg419791

Yes you are right, the 7A24 goes down to 5mV; admittedly I have not used it very much so far, lol. My intention is to collect multiple plug-ins for this scope, so I want a four channel adaptor to make use of, say, a pair or 7A24s.

Those low-tip-c Tek FET probes you mention put the source follower (and input attenuator in the case of the P6202) right in the probe tip. This is a rather different thing from an adaptor intended to permit generic 10:1 10M passive probes to be used with a 50 ohm input.

[GK]

Well it seems to work quite well. Now comes the performance testing. This is a bit of an issue because I only have signal generators that go to 200 MHz and none of my squarewave/pulse generators have particularly fast rise/fall times (well not sub nS anyway).



I rummaged through my parts bins and came up with a 10MHz crystal oscillator module and a 74AC14 hex Schmitt trigger, so I made a real quick and dirty fast rise/fall squarewave source. A 74AC gate will do ~1nS rise and fall with no appreciable capacitive load. So I used the 'AC14 to buffer the oscillator module and decoupled its output from any load capacitance with a 10:1 attenuator. The schematic is shown in the picture below. The attenuator gives me a square wave of 500mVpp amplitude with a ~20 ohm source resistance. I am testing with a Lecroy 500MHz PP006A probe which has 12pF of tip capacitance. 20 ohms corners with 12pF at approx 650MHz. Such high speed logic in a big DIP package is never going to make a nice and clean squarewave with no overshoot, as can be seen in the Rigol scope shot (this is not using the probe adaptor).



Here is what it looks like on my 7904 (on the 500MHz effective BW 7A19 channel) via the PP006A 500MHz bandwidth probe and my 1M probe adapter. The measured rise time on screen is 1nS.  BW=0.35/RT, so it would appear that my board layout for the 1M adapter has achieved a bandwidth of at least 350MHz. How much better I of course cannot measure without a faster rise-time signal source and a higher bandwidth oscilloscope and probe.
 

[GK]

Very nice build!  Please keep us informed 

And it seems to work very well  It will eventually get a write up on my webpage. If anyone is interested I can post up the full schematic and Gerber files before then.

[David Hess]

Those low-tip-c Tek FET probes you mention put the source follower (and input attenuator in the case of the P6202) right in the probe tip. This is a rather different thing from an adaptor intended to permit generic 10:1 10M passive probes to be used with a 50 ohm input.

I just used them as examples of what I was describing.  They were optimized for low input capacitance which is why they lack input protection so even with a 1 megohm input resistance, they would not work anyway unless a lot of shunt capacitance was added.

Incidently with a JFET input, usually only 1 input protection diode is needed to prevent reverse biasing the JFET gate.

I rummaged through my parts bins and came up with a 10MHz crystal oscillator module and a 74AC14 hex Schmitt trigger, so I made a real quick and dirty fast rise/fall squarewave source. A 74AC gate will do ~1nS rise and fall with no appreciable capacitive load. So I used the 'AC14 to buffer the oscillator module and decoupled its output from any load capacitance with a 10:1 attenuator. The schematic is shown in the picture below. The attenuator gives me a square wave of 500mVpp amplitude with a ~20 ohm source resistance. I am testing with a Lecroy 500MHz PP006A probe which has 12pF of tip capacitance. 20 ohms corners with 12pF at approx 650MHz. Such high speed logic in a big DIP package is never going to make a nice and clean squarewave with no overshoot, as can be seen in the Rigol scope shot (this is not using the probe adaptor).

The 74AC14 can do a lot better by flattening it to the ground plane and minimizing lead lengths.  Another thing worth trying is using a small signal schottky diode or NPN base-emitter diode between the 74AC14 output and a 50 ohm parallel termination; then the 74AC14 output disconnects itself on one edge from the output circuit and the highest frequency part of the circuit is minimized.

[GK]

The 74AC14 can do a lot better by flattening it to the ground plane and minimizing lead lengths.  Another thing worth trying is using a small signal schottky diode or NPN base-emitter diode between the 74AC14 output and a 50 ohm parallel termination; then the 74AC14 output disconnects itself on one edge from the output circuit and the highest frequency part of the circuit is minimized.

Yeah the construction could be better (like I said, this was quick and dirty), but with the DIP package not by a whole lot. I was contemplating just what you describe, but with a pair of schottky diodes in anti-parallel and the output of the 74AC14 capacitively coupled to the current limiting resistor to give a +/- squarewave with both polarities clamped. Unfortunately the only suitable low capacitance, very high speed diodes I have immediately at hand can't handle any appreciable current.

[David Hess]

The 74AC14 can do a lot better by flattening it to the ground plane and minimizing lead lengths.  Another thing worth trying is using a small signal schottky diode or NPN base-emitter diode between the 74AC14 output and a 50 ohm parallel termination; then the 74AC14 output disconnects itself on one edge from the output circuit and the highest frequency part of the circuit is minimized.

Yeah the construction could be better (like I said, this was quick and dirty), but with the DIP package not by a whole lot. I was contemplating just what you describe, but with a pair of schottky diodes in anti-parallel and the output of the 74AC14 capacitively coupled to the current limiting resistor to give a +/- squarewave with both polarities clamped. Unfortunately the only suitable low capacitance, very high speed diodes I have immediately at hand can't handle any appreciable current.

The idea is not to clamp the output but to disconnect the driver from the transmission line so that any irregularities in its output are attenuated; the output edge is then determined by the shunt termination and reverse biased diode capacitance which can be very small.  The low capacitance of a diode works well for this and many reference level pulse generators work this way.  Of course only one of the edges is the clean one but usually you only need one for testing.  Through hole discrete implementations can easily achieve 600 picoseconds like this (not fast enough for a 400 MHz test) although I would like to try it using faster transistors and a microwave schottky diode from Avago.

The base-emitter junction of a 2N3904 should work well enough if you do not have any small signal schottky diodes.

Anyway, the above is how I handled transient response calibration before I got a sampling oscilloscope which could be used to verify the performance of a pulse generator.

[BravoV]

If anyone is interested I can post up the full schematic and Gerber files before then.

Please do, thank you for sharing. 

[GK]

If anyone is interested I can post up the full schematic and Gerber files before then.

Please do, thank you for sharing. 

Actually it still needs some work. With the attenuator switched in the frequency response is flat as far as I can measure (to ~200MHz) but I have a high-Q series resonance in the vicinity of 500-600MHz that totally screws up the transient response to fast slewing input signals. This seems to be solvable with the inclusion of some small value series damping resistors but this is going to require a bit of work to get right. I am currently working on a spice simulation of my board layout including track and component parasitic inductances as best as I can estimate them.
   

[GK]

Well this was an interesting experiment. I've spent the whole day at the bench, measuring, modelling and modifying this prototype. In the end I did manage dampen the resonances in the attenuator section to an acceptable level, but only after limiting the bandwidth to 250MHz and eliminating the relays from the signal path. I assessed the performance of the attenuator by comparing the transient response to that of a second OPA653 channel assembled without an input attenuator - ideally, the only difference between the two should be in the signal amplitude, not the transient overshoot or ringing. After that I solder wicked all of the remaining input network components from the board and assembled a passive 10:1 (no relays!) attenuator dead bug style as compact as practical right at the input of the OPA653, with the input BNC relocated to the same location. That worked just as well as the unattenuated reference channel in the full 500MHz bandwidth.

In a nutshell, my initial prototype design would have made an excellent front-end for a 100MHz oscilloscope where all of the =>500MHz infelicities would be masked, but as a front-end for my Tek 7904A, it turned out to be a design failure. So now I find myself ruminating over which alternative direction to take. I still want the 20dB attenuator option for the signal handling capability it provides (think probing the high voltage switching spikes produced by a SMPS), but I can't do without the sensitivity of a 1:1 throughput for small signal stuff either. A selection of AC/DC coupling is also mandatory, but the switching between the two at these frequencies is problematic.

So what I am contemplating doing now is making an eight independent channel design. Four of these channels will have 20dB fixed attenuation and four will have no attenuation. Two of the -20dB channels and two of the 1:1 channels will have fixed DC coupling while the other two channels in each group will have fixed AC coupling. Not quite as convenient as only four channels that can have their mode of operation switched, but far better from an electrical, performance and design perspective as all of the input stage switching (relays) and attendant layout compromises are eliminated. I'll probably house these in a 1U-height 19" rack case.

In addition to all of the above I can report happily that the OPA653 is a fantastic performer. In my entire ~8 hours of experimenting at the bench not once did I witness even the briefest burst of RF oscillation.

[GK]

Looking good.

Yeah, the OPA653 performs well, but my initial prototype passive input stage attenuator not so much. See my post above.

[BravoV]

In a nutshell, my initial prototype design would have made an excellent front-end for a 100MHz oscilloscope ..

That is more than enough for me ... preparing popcorn. 

[joeqsmith]

You know, a 1M scope would solve your problem.    Let me know if I can be of more help.   

[David Hess]

For 200+ MHz designs, Tektronix did not use relays to switch their high impedance attenuators until they were part of a hybrid with one exception and in both cases, they used a slightly different topology.  Instead of using a relay to bypass each attenuator, they used it to short the upper section and disconnect the lower section from ground.

Take a look at how the 7A11 does it using a single SPDT relay for each section.  That may return better results.  The hybrids in the 300+ MHz 2465 series use DPDT relays in a similar way.

[GK]

Let me know if I can be of more help.   

You have a spare Lecroy to donate? 


In hindsight with a bit of modelling the HF limitations of the original design are obvious. There are 2 series resonances in the attenuator that were causing me grief. #1 is the the lower leg compensation capacitor resonating with the net loop inductance. This causes a dip in the frequency response that limits the bandwidth. There isn't much that can be done to eliminate this resonance without screwing up the pass band response even worse. #2 is that of the ~9x lower value compensation capacitor in the upper leg of the attenuator resonating with the net loop inductance. This causes a high-Q, high amplitude, high frequency peaking that rings like a banshee in response to a fast input transient if not damped. There is also a 3rd resonance between the input C of the opamp and the track inductance leading up to it, but I had anticipated this resonance and successfully dealt it with the 100R resistor directly in series with the op-amp input.

#2 is easily damped out of existence with a resistor put in series with the upper attenuator leg (but after the input termination capacitance), but with the penalty of bandwidth reduction as this resistor forms a low pass filter with the input capacitance of the attenuator, which in my initial design was ~6pF. Basically, I couldn't snub this resonance (which was at around 500MHz) out of existence without lowering the bandwidth to ~250MHz or less.

The way to deal with these resonances is two fold. The first is obvious and is to simply minimizing all track lengths and associated inductances. The second is to frequency compensate the attenuator with capacitors that are as small in capacitance value as electrically viable and practical. This shifts all of the pesky resonances to higher up in frequency - preferably so high that they either don't matter or so high that they can be successfully damped out without limiting the bandwidth to an unacceptable level. 

Pictured below is a SPICE simulation of the successfully performing  10:1 attenuator mentioned in my second to last post that I built dead-bug style right at the input of the OPA653. Inductance values for component+track/lead contributions are included. The first sim is without the damping resistors to show the high frequency resonances; the second is with the damping resistances added. I think this is a rough estimate of the limit of performance with 1206 passives. Note that the biggest bugbear is the prominent (bandwidth-limiting) resonance of the 15pF cap causing the dip in the frequency response. If this was only a 5:1 attenuator, this capacitor would only need to be ~half the value, shifting the resonant dip up ~two times higher in frequency. I'm not really comfortable using less than 2pF for the (trimmer) compensation cap in the upper leg, so I think for the -20dB amplifiers I will use 0603 passives for the attenuator. This should net me an attenuator that does not substantially subtract from the performance of the OPA653.

I may have "wasted" a bit of time on an unsuccessful prototype, but now I know for sure how to make a s^%t hot 10:1 frequency-compensated attenuator 

[GK]

For 200+ MHz designs, Tektronix did not use relays to switch their high impedance attenuators until they were part of a hybrid with one exception and in both cases, they used a slightly different topology.  Instead of using a relay to bypass each attenuator, they used it to short the upper section and disconnect the lower section from ground.

Rigol gets away with it to 300MHz (I have the 200MHz version). After some more rumination I think I might have been a little quick to condemn the original design. One thing I didn't try on the original layout before stripping it was to simply reduce the value of the attenuator compensation capacitors to one third of the original design values. As detailed in my previous post that would have shifted the pesky resonances to ~3 times higher in frequency and I could have dealt with the small observed irregularities attributed to the relays simply by increasing the value of the damping resistor in series with the op-amp input to lower the attendant low-pass pole to 300MHz. I think all the original design would require for perfectly acceptable 300MHz performance is a small layout modification to include the additional damping resistor in the attenuator network and the above described changes to a few component values.

I could simply revise the original design along these lines...............hmmmm.............. but now that the bandwidth bug as bit..............

[tggzzz]

For 200+ MHz designs, Tektronix did not use relays to switch their high impedance attenuators until they were part of a hybrid with one exception and in both cases, they used a slightly different topology.  Instead of using a relay to bypass each attenuator, they used it to short the upper section and disconnect the lower section from ground.

This is the front end of a Tek 485, a 50ohm/1Mohm 350MHz scope with useful response to >500MHz.

The BNC input, J1, is at the lower left. To its right are three horizontal sections, the 50ohm attenuator chain, the rotary switch positions, and the high impedance chain.

K1S1 is a relay used to disconnect the internal 50ohm termination (so the input is 1Mohm) when there's an overload.

The switches are exposed flexible fingers on the PCB which are lifted 0.5mm by cams on the rotary switch.

The complete manual is available at various places around the web.