I asked the
HP/Agilent/Keysight interest community on groups.io about the suspcious results I obtained for the phase noise of an FE-5650A using a HP11729C. There were several suggestions why the HP11729C might not provide the correct values, but John Miles (KE5FX), who also contributes to this forum, provided the correct analysis. To ensure this post is self-contained, I repeat the phase noise measurements I obtained:
10 KHz : -124 dBc/Hz
100 KHz: -169 dBc/Hz
180 KHz: -178 dBc/Hz
While the 10 KHz value was in the range of values shown for the FE-5680A (which is supposedly the same electrical device in a different physical package), the value for 100 KHz was much lower than that shown on John Miles's website (see
the phase noise plot). The phase noise graph did not display a value for 180 KHz.
John had used the HP11729C in the past and remembered that for frequencies near the low end of its input range, the input signal sum component in the output of the mixer/LPF drives the LNA into saturation. This means the amplifier cannot properly produce the correct phase noise output.
In the case under consideration, the mixer produces a 20 MHz output component of considerable power, which represents the sum of the two inputs into the mixer (10 MHz). The 15 MHz LPF doesn't sufficiently suppress this sum. In addition, looking at the input to the LNA (which is available as the aux noise output), an even stronger 10 MHz component exists in the mixer output. This component will not be suppressed by the 15 MHz LPF. So, both the 10 MHz and attenuated 20 MHz components drive the LNA into saturation. It isn't clear why the mixer is producing a 10 MHz component, since classically only sum and difference products should appear.
Figure 1 shows the signal output of the 15 MHz LPF (the aux noise output). Notice the significant 10 MHz and 20 MHz frequency components.
Figure 1
Figure 2 shows the output of the LNA. There are significant extraneous spurs visible, which is evidence of LNA saturation. Note: the power of the 10 MHz and 20 MHz components is about 50 dB higher than the aux noise output. The LNA is spec'd at 40 dB gain, so this is additional evidence of the LNA going into compression.
Figure 2
I thank John for this analysis. To be honest, it is not something that would have occurred to me.
To work around this problem, I decided to use the 1Hz-1MHz output of the HP11729C. This output does not use the LNA and has an extra 1.5 MHz LPF that will eliminate the 10 MHz and 20 MHz outputs of the mixer.
The 1Hz-1MHz output has an output impedance of 600 ohm and a voltage range of +/- 10V. Fortunately, I had bought a 600-50 ohm impedance matching pad in case I had to use this output. This pad has an advertised insertion loss of 16.6 dB. Consequently, when processing the raw output, I included a correction that added 16.6 dB to each data point (in addition to the other corrections specified in
a previous post). Since a voltage swing of +/- 10V represents a signal with maximum power of 30 dBm, with the 600-50 ohm matching pad in place, the maximum power of this signal would be 13.4 dBm. On the other hand, since the two inputs are in quadrature, it is likely the actual voltage swing of the output will be much less.
I ran the phase noise experiment. The processed spectrum (10Hz-200Hz) is shown in figure 3.
Figure 3
This yielded better results for 100Khz. Specifically,
10 KHz : -96 dBc/Hz
100 KHz: -122 dBc/Hz
Comparing these results with those on John Miles website reveals some puzzles. In particular, the published results are:
10 KHz: lower bound of around -125 dBc/Hz and upper bound of around -100 dBc/Hz (the data is in a graph and not presented numerically)
100 KHz: lower bound of around -133 dbc/Hz and upper bound of around -113 dBc/Hz
The 100 KHz result from the HP11729C is within the bounds of the published results, but the HP11729C 10 KHz result is about 4 dB higher than the upper bound of the published result.
I decided to back out the 16.6 dB correction for the impedance matching pad and see what happened. Figure 4 shows the result.
Figure 4
Numerically, the offsets of interest are:
10 KHz : -112 dBc/Hz
100 KHz: -138 dBc/Hz
Now the 10 KHz value is within the range of the published result, while the 100 KHz result is about 5 dB lower than the published result's lower bound.
What to do? After stewing on this for a while, I had an idea. Suppose the published figure for the impedance matching pad insertion loss was too high? If it was somewhat lower, both the 10 KHz and 100 KHz experimental results might conform to the published result.
So, I measured the insertion loss of the impedance matching pad. This was a bit tricky, since the only device I have with connectors at 600 ohm output impedance is the HP11729C. Furthermore, I couldn't use an input signal that would be filtered by the 1.5 MHz LPF in front of the 1Hz-1MHz output.
To begin with, I purchased a 600 ohm BNC terminator, which arrived last weeked. Needing only some signal of sufficient power coming from the 1Hz-1MHz output, I did the following. I input a 1 MHz/200 mV signal from my DG1022 to the 5-1028 MHz input (one of the mixer inputs) of the HP11729C. I then input a second 1 MHz/200 mV signal phase shifted by 90 degrees into the Microwave test signal input (the other mixer input). I connected the 1Hz-1MHz output to my scope using a 3' coax (which at 1 MHz should not have transmission line characteristics), first through a BNC-T terminated with the 600 ohm terminator and then through the impedance matching pad to the BNC-T terminated with a 50 ohm terminator.
Figure 5 shows the result without the impedance matching pad and Figure 6 shows the result with the pad.
Figure 5
Figure 6
The (rough) peak-to-peak voltages are 9 mV or -36.9 dBm and 1.1 mV or -55.19 dBm. This is a difference of 18.29 dB. The measurements on my scope (a Rigol 1104Z) using a crude cursor set up at the lower limit of the scope's voltage range are not definitive. Nevertheless, it doesn't seem like the insertion loss is less than 16.6 dB. So, this eliminated the hypothesis I was considering.
I then considered the possibility that the phase noise data published on John Miles's website was averaged over a significant interval of time. This seems reasonable, since the FE-5650A is intended as a component in a time-keeping device.
The sweep interval selected by my SA for 10KHz-200KHz at 10 Hz RBW was about 17 seconds. So, I decided to lengthen this interval to see if that brought the experimental data into line with the published results.
Unfortunately, the Siglent SSA3032X would not let me increase the sweep interval when the RBW is 10 Hz. I had to increase RBW to 1 KHz to execute this experiment.
Figure 7 shows the results from a 6 second sweep, whereas Figure 8 shows the results from a 300 second sweep.
Figure 7
Figure 8
A cursory examination of the plots shows that increasing the sweep interval actually increased the measured phase noise. For example, the (raw and uncorrected) value for 10 KHz from the 6 second sweep is ~ -49 dBm, whereas the (raw and uncorrected) value for 10 KHz from the 300 second sweep is ~ -44 dBm. In other words measured phase noise gets worse for longer averging times.
At this point, I decided to present the results I have so far obtained and ask for comments. As things stand now, I can think of 4 possibilities for the discrepancies between the experimental results I obtained and those published on John Miles's website:
• There is a problem with my measurement methodolgoy or its execution. |
• There is a problem with the published results. |
• The FE-5680A and FE-5650A are not identical except for packaging. The published results do not apply to the FE-5650A. |
• The published results are for a freshly minted FE-5680A, whereas my experimental results are for a 15 year-old FE-5650A. Aging has deteriorated the phase noise performance of the latter. |
I would be interested in comments addressing these possible explanations or other explanations for the discrepencies between the results I have obtained and the published results.